Digital resampling method and apparatus

ABSTRACT

A method for digital resampling in a digital communications receiver is described. The method comprises selecting whole samples of a received input signal in the time domain and implementing sub-sample digital interpolation in the frequency domain. This amounts to performing the time shift of the interpolation process in the frequency domain. The method may be performed in conjunction with the operation of a polarisation recovery filter. A digital communications receiver is also provided the receiver being arranged to perform frequency domain sub-sample interpolation on an input data signal.

PRIORITY CLAIM

This application claims priority to United Kingdom Patent ApplicationNo. 1819215.3, filed Nov. 26, 2018, the entirety of which isincorporated herein by reference.

TECHNICAL FIELD

The present disclosure relates generally to a method and apparatus forperforming digital resampling in an optical communications system.

BACKGROUND

Modern optical communications systems now routinely employ coherentdetection and digital signal processing in the transmission of dataalong optical signal paths between transmission and receiver devices. Insuch systems, information is encoded into the modulated amplitude,phase, and polarisation of an optical signal, achieving very highoptical channel capacity compared with systems using, for example,on-off keying only. Optical signals comprise an X polarised componentand a Y polarised component.

Quadrature amplitude modulation (QAM) is a modulation method in whichthe optical waveform transmitted through an optical channel comprises acombination of phase-shift keying (PSK) and amplitude-shift keying(ASK). Higher order QAM modulation schemes (64QAM or 128QAM) arebecoming increasingly common as signal processing becomes more efficientand data rates increase. However, the faster data rates and increasedspectral efficiency provided by the higher order modulation techniquesare by their nature more susceptible to noise and interference.

Optical signals become distorted during their transmission along anoptical signal path. In the case of optical fibres, imperfections in thesurface of the fibre, or asymmetries in the cross-section of the fibrewill cause dispersion in the polarisation mode (known as PMD) orpolarisation dependent loss (PDL). Different path lengths inside thefibre will cause time related distortion effects. The materialproperties of the fibre can lead to chromatic dispersion (CD).

Polarisation mode dispersion (PMD) is a result of an optical pulsepropagating through a fibre in two randomly coupled polarisation modes,and varies with optical frequency.

Signals are transmitted along an optical fibre with reference to a clocksignal. The received signal will be subject to jitter and timingfrequency uncertainty, and so clock recovery processing is used.

When processing a signal at a receiver device in a system employing QAM,digital resampling of the received input signal is often performed.Digital resampling uses interpolation filters to reconstruct a discreterepresentation the original signal at a new sample rate. Finite impulseresponse (FIR) filters may be used in order to remove jitter effects soas to accurately resample the signal. In order for a FIR filter to moreeffectively remove higher frequency jitter effects introduced in thetransmission channel, so that the signal can be accurately re-created, alonger (or higher order) filter is used. FIR filters introduce delay andmultiplication to the signal path in order to create the filteringeffects. A longer filter leads therefore to more calculations on thesignal per clock cycle, which increases both power demands and latencyin the signal path.

A butterfly structure of FIR filters may be used when polarisation of asignal is in an unknown state.

As can be appreciated, the longer/higher order a filter, the moreintense the calculation in operation. It may be generally stated that asthe complexity of the modulation of a signal increases, so too does thecomplexity of re-creating the original signal at the receiver, and wherecomputational complexity increases, so too do the power demands upon asystem.

BRIEF DESCRIPTION OF THE DRAWINGS

Examples of the present disclosure will now be explained with referenceto the accompanying drawings in which:

FIG. 1 is a block diagram of a prior art coherent optical communicationsreceiver comprising means for clock recovery;

FIG. 2 is a block diagram of an optical communications receiver asdescribed herein;

FIG. 3 is a flow chart describing a method for digital resampling asdescribed herein;

FIG. 4 is a block diagram illustrating a digital resampler and clockrecovery module for performing the methods described herein;

FIG. 5 is a block diagram of a sample selector for use as part of themethod and digital resampler described herein

FIG. 6 is a block diagram illustrating digital resampler combined with adynamic filter module, and clock recovery module as described herein;

FIG. 7 is a block diagram of part of a combined polarisation filter andresampler as described herein;

FIG. 8 is a block diagram of a combined polarisation filter andresampler as described herein; and

FIG. 9 is a flow chart describing a method for combined digitalresampling and dynamic filtering of an input signal as described herein.

Throughout the description and the drawings, like reference numeralsrefer to like parts.

DESCRIPTION OF EXAMPLE EMBODIMENTS Overview

A method for digital resampling in a digital communications receiver isdescribed herein. The method comprises selecting whole samples of areceived input signal in the time domain and implementing sub-sampledigital interpolation in the frequency domain. This amounts toperforming the time shift of the interpolation process in the frequencydomain. The method may be performed in conjunction with the operation ofa polarisation recovery filter. A digital communications receiver isalso described herein, the receiver being arranged to perform frequencydomain sub-sample interpolation on an input data signal. Optionalfeatures are set out in the appended dependent claims.

EXAMPLE EMBODIMENTS

FIG. 1 is a block diagram of an example digital signal processor (DSP)module 100 of the prior art for use in an optical communicationsreceiver modem (not shown). A modem receiving an optical signalcomprises a transceiver for converting the signal to an electricalsignal, and this is passed to an analogue-to-digital (ADC) converter,which takes the analogue data stream signal and outputs a discretedigital signal having a particular sample rate. The ADC output samplesare affected by the distortions and noise introduced in the opticaltransmission system and so the signal is processed and resampled inorder to recreate the original transmitted signal and eliminate dataerrors. The ADC 102 may comprise four distinct ADCs in the case of dualpolarisation QAM, one each to treat the real and imaginary components ofthe X and Y polarisation components present in the received signal (XI,XQ, YI and YQ). The output of the ADCs may be passed to a chromaticdispersion (CD) equaliser module 104, which is arranged to treat CD.Once CD has been removed, the signal is passed to polarisation equalisermodule 106, and subsequently to carrier recovery module 108.

Timing error detector 110 is provided to detect errors in the samplerate of the ADC with respect to the rate at which the signal wasoriginally transmitted. Timing error detector 110 may receive as itsinput the outputs of ADC 102, CD equaliser module 104, polarisationequaliser module 106, and carrier recovery module 108. Timing errordetector is arranged to feed back to a loop filter 120, which in turncontrols a voltage controlled oscillator (VCO) 130. VCO 130 provides thereference clock signal to the ADC 102 which determines the nominalreceive frequency based on the timing error determined by timing errordetector 110. The feedback loop shown in FIG. 1 provides a way to adjusta sampling reference clock rate at the ADC so as to minimise samplingerrors of the incoming digital signal.

If a timing error detector takes the output of the ADC 102 as its input,system latency in the DSP of FIG. 1 is clearly minimised, since thesignal is received directly from the ADC without any further processingacting on the signal. However, the distortion and jitter still presentin the signal at this point make control of the VCO 130 very difficultdue to the level of distortion on the signal. It will be appreciatedthat as the CD equaliser module 104, polarisation equaliser module 106,and carrier recovery module 108 act on the input signal, the signalprovided to the input of the timing error detector will be “cleaner”with respect to the distortion errors in turn. As more latency isintroduced to the timing error detector 110 and VCO 130 signal path bythe processing of the modules shown, the available loop filter 120bandwidth is reduced, and therefore the ability to compensate for higherfrequency jitter is reduced. When filtering jitter from a signalmodulated using higher order modulation schemes, high accuracy isrequired, and so the longer the filter the better. However, as shown,there is a trade-off between quality of filtering and latency introducedinto the signal path.

Low-power digital resampling capable of treating signals having higherorder modulation, and therefore higher frequency jitter, is required.

FIG. 2 shows a block diagram of an architecture of a DSP 200. DSP 200comprises an ADC module 202, a signal preparation module 204, resamplermodule 206, and clock recovery module 208. ADC 202 is arranged toreceive an input high speed data signal which may comprisedual-polarised (X, Y) signal data. Where dual polarised signal data isthe input to the modem, input data is separated in two complex signals,one for the X polarization, and one for the Y polarisation. The complexvalued signals can again be separated in two real-valued signals I andQ. (In phase and Quadrature component, often also called 0- and90-degree component).

The signal processing module 204 prepares the digital samples forfurther processing. Signal scaling and chromatic dispersioncompensation, among other functions, may be performed here.

The resampler module 206 performs fast jitter removal and frequencyrecovery for modest frequency offsets. Resampler module 206 alsocomprises a polarisation filter. The resampler module 206 is only ableto introduce a timing phase shift, such that frequency offsets arecompensated by changing the ADC clock frequency. The resampler module206 also contains a digital phase-lock-loop (PLL) that uses a phasedetector signal from clock recovery module 208 to track clock jitter.The control signals for the interpolation carried out in the resamplermodule 206 are derived from the clock recovery module.

Generic resampler function will now be described. Data is sampled atfrequency f_(x) by the ADC and put into a buffer. Data is also extractedby the interpolator at the rate f_(k). Input and Output pointers aregenerated by a pointer calculator in clock recovery module 208, and areincremented at their respective rates. The output lags behind the input.Whilst the input pointer is an integer, the output pointer can benon-integer since interpolation will be performed in the resampler.

Mathematically we can describe the behaviour as follows. The data isregularly (frequency f_(x)) placed into the buffer at integer pointerp_(in):

${p_{i\; n}(t)} = \frac{t}{T_{x}}$

Data is regularly (frequency f_(k)) extracted from the buffer atnon-integer point p_(out):

${p_{out}(t)} = {\frac{t}{T_{k\;}} - L}$

where L is the time lag in the buffer to ensure that input and outputpointers do not collide. At any output index k, the output index isreading from pointer:

p _(out) =γk

and the input index is writing (synchronously or asynchronously) topointer

p _(in) =x=└γk┘+L

where γ is the ratio between input and output sample rates.

Resampling is significantly simplified when the input and output samplerates are almost identical. Here γ has the value ≈1.0. In the followingselections, the ratio γ is replaced by the error signal β=γ−1.

Note that we are referring to sample rates rather than clock rates. Theratio γ is independent upon the degree of parallelism at the input andoutput. The clock rates are managed so that input and output pointers donot collide. This is a separate task handled by a Pointer DifferenceCalculator (not shown).

The clock recovery module 208 comprises a phase detector which controlsthe resampler module 206 via a digital loop filter. A pointer calculatorand VCO loop-filter are used to control the ADC reference clock.

Control signals (integer and fractional pointers) are generated in theclock recovery module 208 and used to control the resampler module.Further function of modules 206 and 208 will be described in detailbelow.

DSP 200 may further comprise carrier phase estimation module 210 andforward error correction module 212, which provide the signal toserialiser/deserialiser 214 for subsequent onward electricaltransmission.

FIG. 3 shows a flowchart describing a method for digital resampling in acoherent optical communications receiver. At step 302, data samples arereceived. The data samples have been converted from an optical datainput stream into a sampled data input stream comprising a plurality ofdata samples having a first sample rate. The sampled data input streammay have been treated by the signal pre-processing steps described abovein relation to FIG. 2.

At step 304, whole samples are selected from the digital input signal inthe time domain. The selection of the whole samples is determined by aninteger pointer value which may be received from a pointer calculator. Asample selector receives, for example, N samples per clock cycle as aninput, and the samples are placed into a buffer. The integer pointervalue, or base pointer, p₀(t), defines the first sample from the bufferwhich is to be extracted. The selected samples comprise at least as manysamples as those received per clock cycle (at least N in this example),as will be explained further below.

At step 306 complex time shift values are generated for the selectedwhole data samples. The complex time shift values are generated based ona fractional pointer value received from a pointer calculator. Thegeneration of the complex time shift values may comprise providing thefractional pointer value as an input to a lookup table. The pointer,comprising the integer and fractional parts, may be derived from theoutput of a digital loop filter, as will be further explained below.

Sub-sample interpolation is then performed on the selected samples byapplying the complex time shift values to the selected whole samples(308). Applying the time shift values is done by multiplying therespective values by the selected samples, and this takes place in thefrequency domain.

FIG. 4 shows a block diagram of a resampler and clock recovery module408, for performing a digital resampling method according to the presentdisclosure.

A resampler module 406 receives an input data signal as part of thedigital signal processing at an optical data receiver modem. The inputdata signal is passed to a sample selector buffer 412. The length of thebuffer may be determined by the amount of jitter which may affect theinput data signal. As will be appreciated by the skilled person, higherorder modulation formats leads to the use of longer interpolationfilters to minimise any added distortion. As will be described furtherbelow, the number of samples extracted from the buffer may depend on afilter length, with a higher order filter leading to more samples beingextracted. It will also depend upon the number of samples to beprocessed in a single step, using a frequency domain filter.

Sample selector 414 selects a predetermined number of samples from thesample selector buffer 412, using the integer part of the pointer(termed base pointer) and provides them to interpolator 416. Theinterpolator uses these samples in conjunction with the respectivefractional part of the pointer (fractional pointer).

The resampler module 406 works in conjunction with clock recovery module408. The clock recovery module 408 contains a timing phase detectorwhich analyses the data, and detects any timing phase error. This erroris used with a loop filter to adjust the desired timing phase—where the“Pointer Calculator” 426 reflects this by changing the output pointerwith integer and fractional parts.

The particular selected samples which are to be taken from the sampleselector buffer 412 are determined by the integer pointer value receivedat the sample selector from the pointer calculator. The integer pointervalue, p₀(t), may define the first sample in the buffer which is to beselected, of the predetermined number of samples. These samples areselected in the time domain. The integer pointer value p₀(t) may be usedto define another sample of the samples to be selected, such as themiddle sample, or the final sample.

Interpolator 416 receives the selected whole samples from the sampleselector 414, and converts them into the frequency domain. Theinterpolator receives the fractional part of the pointer from thepointer calculator 426, and generates complex time shift values for theselected samples. An example form which the shift values may take isthat of a lookup table, having a predetermined number of values. In theexample case where 64 values are to be generated, the following equationdetermines the complex time-shift values:

${\exp\left( {j\; 2{{pi} \cdot \mu \cdot \frac{\left\lbrack {{0\text{:}31},{{{- 32}\text{:}} - 1}} \right\rbrack}{64}}} \right)},$

where μ is the fractional pointer value. Other techniques such as Cordicalgorithms are also possible but incur more latency.

The interpolator 416 multiplies the complex time shift values with therespective frequency domain selected samples in order to causesub-sample interpolation. This creates a set of interpolated signalsamples, which may be further treated accordingly.

Performing interpolation in the frequency domain using complex timeshift values derived from a fractional pointer is a more efficientmethod of interpolation than applying FIR filters in the time domain.This is somewhat counter-intuitive, because converting the signal intothe frequency domain results in a processing of the signal, which placespower demands in the signal path. However, frequency domain filtering isknown to be efficient for block-wise processing, especially with filterlength greater than 5 or 6 tap weights. In addition, as explainedfurther below, the frequency domain interpolation may beopportunistically combined with an existing frequency filter forcompensating polarisation effects.

The conversion of the selected samples into the frequency domain may bedone using a Fast Fourier Transform (FFT) function. The FFT has a lengthN, and the output of the FFT (an FFT output) is multiplied by the Ncomplex time shift values (complex valued numbers); that is to say, ashift generate module therefore generates one complex shift value foreach frequency of the FFT).

In order to continue processing the signal in the time domain, aninverse FFT (IFFT) may be performed on the samples.

FIG. 5 shows a sample selector module 500 for use with the methodsdescribed herein. The sample selector module 500 comprises a sample bus512 and a sample selector 514. Sample selector module 500 receives aninput signal at a first sample rate. The first sample rate may be higherthan that of the signal received by the optical receiver modem, in orderto allow for compensation for channel distortions of the signal and toprovide the bandwidth that will be used for interpolation. For example,the input signal be sampled at a sample rate of 4/3 samples/signal. Thereceiver as a whole may operate on a clock cycle of the sample ratedivided by a fixed number of samples—for example 256 resulting in 256samples being processed in parallel per clock cycle.

At each clock cycle interval, 256 new samples from the input data streamare placed in the buffer, in block 2. The previous clock cycle's (Z⁻¹)samples are moved to block 1. Longer buffers will result in severalclock cycles being stored in the buffer (subsequent Z⁻¹ in FIG. 5). Asmentioned above, the skilled person will appreciate that the samplebuffer may be adapted to suit the system in which it is provided, andthe values presented here are for illustration purposes. A longer orshorter buffer may be used within the scope of the present disclosure.

The sample selector 514 receives input pointer p₀(t) from the pointercalculator. The p₀(t) value received defines the first sample to betaken from the buffer. The sample selector 514 then takes apredetermined number of samples from the buffer using the defined firstsample. Where the clock rate is sample rate/N, N samples may be taken.If further processing (for example filtering—described below) is to becarried out, N+K samples may be taken, where K is a predetermined numberdepending on the type of further processing or filtering to be carriedout on the signal. The selected samples are then passed to aninterpolation module for frequency domain sub sample interpolation.

If the system is functioning correctly, and the clock is operating withno offset, then the sample selector remains in the middle of the buffer.Where there is a frequency offset between the clock and the inputsignal, we would expect the sample selector to drift away from thecentre. Where, for example, a sinusoidal jitter is present on the inputsignal, we would expect a similar sinusoidal effect on the sampleselector as the clock rate lags or anticipates the jitter. It will beappreciated that the number of clock cycles' data which is stored in thebuffer could include more than that shown in FIG. 5; for example, if onewanted to compensate for larger amounts of jitter, then a larger bufferwould be used.

The frequency-based interpolation method described herein may becombined with upsampling where appropriate in the context of the opticalreceiver modem.

FIG. 6 shows a simplified block diagram of a combined resampler anddigital polarisation recovery filter 610 and clock recovery module 620.The polarisation-filter-plus-resampler is similar to that of theresampler of FIG. 4, but includes a polarisation filteringMultiple-input-multiple output (MIMO) filter module 618. The dashed linearound interpolator 616 and filter module 618 indicates that the twofunctions may be combined. The interpolation carried out by interpolator616 may take place in the frequency domain within the filter module.Sample selector 614 selects a predetermined number of samples fromsample selector buffer 612. The interpolator 616 receives selectedsamples from sample selector 614 and passes retimed data 617 through thefilter module 618 which performs the dynamic filter functionality. Thisis described in more detail below.

The combined resampler and digital recovery filter 610 receives integerpointer p₀(t) and fractional pointer μ₀(t) from pointer calculator 626.Clock recovery module 620 comprises timing phase detector 622 and adigital loop filter 624. The timing phase detector 622 determines atiming phase error of the data at the output of the polarisationfilter+resampler, which ideally is zero. The phase information isextracted from the different data streams (XI, XQ, YI, YQ).

The pointer calculator 626 generates the integer and fractional outputpointers to remove the timing phase error. A common fractional pointerμ₀(t) may be used for all interpolators.

The pointer calculator 626 is also used to generate a phase detectorvalue that can be used to control the ADC VCO. As the output pointermoves away from the centre of the sample selector buffer 612, thePointer Calculator compensates by changing the rate of the receiver'sdata receive clock (ADC ref clock), via a VCO loop filter (not shown).

The phase and frequency of the resampling carried out is dependent upona resampling ratio γ. In an example, there is a modest frequency offsetto be compensated such that γ≈1 and the residual error β=γ≈10. A digitalloop filter 624, containing proportional and integral filters, takes thecurrent phase error from the timing phase detector, and generates acontrol signal for the pointer calculator such that the phase error willbe reduced.

The dynamic filtering carried out in the resampler shown in FIG. 6 mayincorporate an overlap and save method. The overlap and save method is afrequency domain implementation of the FIR filters which are describedabove. The filter will be implemented in the frequency domain using theoverlap and save method. A block of input data is concatenated with thelast input block and converted to the frequency domain (FFT). The datais then multiplied by frequency domain taps H, which compensate forpolarization effects and perform matched filtering. H is a matrix ofcomplex valued tap-weight vectors received from a tap-update moduledelivered from a tap update module, where

$\overset{\_}{H} = \begin{pmatrix}{\overset{\_}{H}{xx}} & {\overset{\_}{H}{xy}} \\{\overset{\_}{H}{yx}} & {\overset{\_}{H}{yy}}\end{pmatrix}$

where H_(xx), H_(yx) are the taps for the x polarization, and H_(xy),H_(yy) represent the taps for the y-polarisation.

The data is then converted back to the time-domain (typically usingIFFT). Finally, the first part of the output block is discarded.

In both frequency and time-domain implementations care must be taken toalign input data and any error signal.

If we wish to implement the equivalent of an N-tap FIR filter, that isto say, an FIR filter with N coefficients, in the frequency domain, wetake the samples for clock cycle (e.g. 256 samples) plus N−1 additionalsamples. An FFT is then performed on these samples, which are thenmultiplied by frequency domain tap weight values. An inverse Fouriertransform (IFFT) is then performed to bring the signal back into thetime domain. The N−1 samples are then discarded from the output of theIFFT, and the output is concatenated with the data from the previousclock cycle, to produce the filtered output signal.

The sample selection for the filter modes described may be achievedusing the sample selector shown in FIG. 5. In an example case, thesample selector 514 has a length 256+K samples, where K is equal to thenumber of taps in the filter minus 1. In the case where, for example, a21 tap filter is implemented, sample selector 514 outputs 276 samples(256+20). The buffer itself is significantly longer to take account oftiming jitter that is to be compensated. It will be apparent to theskilled person that the foregoing figures are modifiable according tothe requirements and specification of the receiver modem or other systemin which the sample selector is implemented.

FIG. 7 shows a combined polarisation filter and resampler (DFIL) 700.Input data is received at DFIL 700, and is passed to sample selector702. The input data may comprise X and Y polarised Data [I, Q], at afirst sample rate, for example 256 samples/clk. Sample selector 702 alsoreceives the integer part of a pointer value received from a pointercalculator (not shown) as described in the foregoing. The sampleselector 702 operates in the way described above with reference to FIGS.5 and 6. The selected samples, based on the integer pointer value p₀(t),are passed to the “data sort-in” module 706.

Generate shifts module 704 receives the fractional part of the pointervalue from the pointer calculator. Generate shifts module 704 isarranged to output a set of complex time shift values in the form s(0, .. . N−1). This may be in the form of a Lookup table (LUT). The lookuptable can effectively be hardwired into the circuitry of the generateshifts module. The output from the generate shifts module can changeevery clock cycle, since it is dependent on the fractional pointervalue, which may be updated every clock cycle.

The base pointer and fractional pointer are treated coherently—that isthey are the integer and fractional parts of a single variable. Thefractional pointer may be buffered before the generate shifts module 704so as to account for any latency in the FFT and sample selector modules.

Data sort-in module 706 sorts the samples and distributes them forconversion into the frequency domain. The output of data sorting moduleis provided to one or more FFT modules (708 a, 708 b, . . . ). Thenumber of signal paths depends on the requirements of the system. Anexample is given below. The FFT module(s) 708 a/708 b produces an outputwith length N (i.e. the same length as the lookup table of complex timeshift values). The output of the FFT is a data array of the form d(0, .. . N−1).

The complex time shift values are applied to the output of the FFTmodules 708 a and 708 b at time shift modules 710 a and 710 b. Thetime-shift block is a set of complex multiplications.

The output of the time shift module is then passed to the polarisationfilter modules (POLs) 712 a and 712 b. The POL(s) 712 a/712 b is (are)used to compensate for time-varying and polarisation dependent effectssuch as PMD etc. described above. It also removes residual chromaticdispersion and acts as a matched filter.

The sample-selector module and digital interpolator associated with theresampler are also implemented within the polarisation filter, thesample-selector being performed in the time-domain, and theinterpolation and polarisation filtering together in the frequencydomain.

The POL(s) 712 a/712 b may include a MIMO filter consisting of 4 complexvalued, frequency domain filters arranged in a butterfly structure ascan be seen in FIG. 8. FIG. 8 shows an abstraction of the combined POL712 and resampler according to the present disclosure. In overview, Xand Y polarized data, Data-X and Data-Y are received, and the functionalprocesses outlined above are performed on each of them. In POL 712, Xand Y frequency domain data are exchanged and multiplied by thefrequency domain compensation response Hx=[Hxx Hxy] for the xpolarization filter and Hy=[Hyy Hyx] for the y polarization filter, suchthat the result is

${\left\lbrack {{Hxx}\mspace{14mu} {Hyx}} \right\rbrack \begin{bmatrix}X \\Y\end{bmatrix}}\mspace{14mu} {{{and}\mspace{14mu}\left\lbrack {{Hyy}\mspace{14mu} {Hxy}} \right\rbrack}\begin{bmatrix}Y \\X\end{bmatrix}}$

respectively. In the presented implementation, this is performed in oneor more sub-filters, each with 64 frequency points.

At the output of the POL(s) 712 a/712 b the data is converted back intothe time domain via an inverse FFT (IFFT) module(s) 716 a/716 b. Padmodules 714 a and 714 b are provided. The pad module(s) 714 a/714 binsert(s) zero values at the centre of the IFFT function. The length ofthe IFFT is different to that of the FFT, and in this way the inputsample rate may be upsampled (or downsampled) to provide the resamplefunction. For example, an FFT-64 and IFFT-96 may be used, which providesa 3/2× upsampling. In an example receiver having an input sample rate of4/3 samples/symbol, the combined polarisation filter-plus-resampler andinterpolator outputs at 2 samples/symbol.

The time-domain output of the IFFT module(s) 716 a, 716 b is passed to“data sort-out” module 718 and output to the clock recovery module. Thesample rate of the input data to the clock recovery module is differentto the sample rate of the data at the input of theresampler-polarisation filter and timing error has been corrected.

DFIL 700, 800 comprises parallel sub-filter paths (a) (b) and mayinclude as many as required by the system requirements outlined in theforegoing. In an exemplary embodiment 6 parallel sub-filter paths may beprovided and the POL+resampler may operate in a 21 tap mode.

As described above, the samples selected by sample selector aredistributed between the FFT filters and resorted at the output.

The following table shows an example distribution of data samples to thesub-filter signal paths where six sub-filters are used.

TABLE 1 Bus mapping Bus Extension K = 20 Input:: Input:: IFFT:: IFFT::Output:: Output:: FFT no. start stop IFFT no. start stop start stopFFT64 [0] 0 63 IFFT96[0] 15 80 0 65 FFT64 [1] 44 107 IFFT96[1] 15 80 66131 FFT64 [2] 88 151 IFFT96[2] 15 80 132 197 FFT64 [3] 132 195 IFFT96[3]15 80 198 263 FFT64 [4] 176 239 IFFT96[4] 15 80 264 329 FFT64 [5] 212275 IFFT96[5] 27 80 330 383

It can be seen that the data sorting module provides the sample overlap(20 samples) used to implement an overlap and save method used by thePOL+resampler in a 21 tap mode.

The complex shift values are applied to the samples in each of theactive sub-filter paths, as outlined in the foregoing; the same complexshift values may be applied to each of the sub-filter paths. Thefractional pointer, μ0(t) value may have 7 input bits in order that thesystem is able to remove jitter at the order of 0.01 UI—where 1 UI isthe width between adjacent eye openings on a conventional “eye diagram”of the input signal.

The foregoing describes a combined POL+resampler. This combinationprovides a more efficient method for removing jitter in a signalprocessed by a coherent optical communications receiver. By performingthe interpolation function in the frequency domain, an increase in thelength of filter is achieved, for the same amount of processing.

The 21 tap mode described above, with 6 sub filters operating throughFFT modules having length 64, uses 3072 real multipliers to perform theinterpolation on two polarisations of the X,Y polarised signal. A knowntime-domain based FIR interpolator system having 6 taps uses the samecomplexity of system (i.e. 3072 multipliers in the 6-tap filteringprocess). This means that the frequency domain-based interpolationcarried out by the present system achieves greater precision in jitterremoval, for the same processing power used by a time domain basedinterpolation having fewer taps.

The present system also uses existing components in an opticalcommunications receiver, in the DFIL implementation which already usesan FFT/IFFT combination to perform the dynamic filtering, therebyachieving further efficiency without adding further signal processingand therefore latency into the signal path.

FIG. 9 describes the steps of a method for digital resampling in acoherent optical communications receiver. The digital resampling takesplace in a Dynamic polarisation recovery filter module (DFIL) andcombines sub sample interpolation with the features of the DFIL'sdynamic filter functions. The steps in the figure comprising dashedlines are part of the function of the optical communications receiver,and are not necessarily part of the method. They are provided forbackground and context. At step 902 an optical signal is received at thereceiver. At step 904 the optical signal is converted to an electricalsignal at a transceiver module provided at the receiver. The receivedoptical signal may be a modulated signal according to a known modulationscheme, such as those described above. The electrical signal isconverted to a digital signal having a first sample rate at step 906.This is performed by an ADC. Prefiltering of the signal may take placeat step 907. At step 908, whole samples are selected from the digitalinput signal in the time domain. The samples are selected based on aninteger pointer which is generated by a pointer calculator. The numberof whole samples selected may be dependent on an internal clock rate,the amount of distortion in the input signal (and thereforeamount/frequency of jitter to be removed) and the modulation scheme ofthe input signal. Complex time shift values are generated at step 910.The complex shift values are generated based on the fractional part ofthe pointer generated by the pointer calculator. The selected wholesamples are divided among one or more sub-filter signal paths at step912. The dividing of the samples is done such that each sub filter pathcomprises samples present in another sub-filter signal path, in orderthat filtering according to the overlap-and save method may be used.Steps 914 to 920 are applied to each of the sub-filter signal paths. Atstep 914 an FFT is performed on the samples in each respectivesub-filter signal path to convert them to the frequency domain. Thecomplex time shift values are applied to the frequency domain samples atstep 916, and this provides sub sample interpolation of the signal,being a time shift in the frequency domain. The time shifted samples arethen filtered at step 918 using the feedback loops described above. Thefiltered signals are passed through an IFFT at step 920 to bring themback into the time domain. The IFFT has a different length to that ofthe FFT. This can be achieved by padding a certain amount of the centreof the IFFT with zeros—for example, where an FFT-64 is used, and IFFT-96can be used, with the centre 32 samples set to zero. This modifies thesample rate of the signal to a second sample rate. At 922 the data fromthe sub-filters is recompiled.

For ease of explanation, the steps of FIG. 9 have been indicated asbeing performed sequentially. However, as the skilled person wouldunderstand, these steps may be performed continuously, in paralleland/or in real-time.

An effect of the present disclosure is the performing of more complexfiltering of an input signal than in previous coherent opticalcommunications receiver modems, without the associated and expectedincrease in signal processing.

It should be understood that, in the present disclosure, the verb ‘tooperate’ may be used to mean ‘to execute’ or ‘to run’.

Those skilled in the art will recognise that a wide variety ofmodifications, alterations, and combinations can be made with respect tothe above described examples without departing from the scope of thedisclosed concepts, and that such modifications, alterations, andcombinations are to be viewed as being within the ambit of the disclosedconcepts.

Those skilled in the art will also recognise that the scope of theinvention is not limited by the examples described herein, but isinstead defined by the appended claims.

The approaches described herein may be embodied on a computer-readablemedium, which may be a non-transitory computer-readable medium. Thecomputer-readable medium carries computer-readable instructions arrangedfor execution upon a processor so as to make the processor carry out anyor all of the methods described herein.

The term “computer-readable medium” as used herein refers to any mediumthat stores data and/or instructions for causing a processor to operatein a specific manner. Such storage medium may comprise non-volatilemedia and/or volatile media. Non-volatile media may include, forexample, optical or magnetic disks. Volatile media may include dynamicmemory. Exemplary forms of storage medium include, a floppy disk, aflexible disk, a hard disk, a solid state drive, a magnetic tape, or anyother magnetic data storage medium, a CD-ROM, any other optical datastorage medium, any physical medium with one or more patterns of holes,a RAM, a PROM, an EPROM, a FLASH-EPROM, NVRAM, and any other memory chipor cartridge.

In one form, a method for digital resampling in a coherent opticalcommunications receiver, the coherent optical communications receiverbeing adapted to convert an optical data input stream into a digitalinput signal comprising a plurality of data samples at a first samplerate, is provided. The method comprises: selecting whole samples fromthe digital input signal in a time domain based upon an integer pointervalue, to provide selected whole samples; generating complex time shiftvalues for the selected whole samples based upon a fractional pointervalue; and creating interpolated signal samples by applying the complextime shift values in a frequency domain to the selected whole samples toperform a sub-sample interpolation of the selected whole samples.

In one example, generating the complex time shift values comprisesproviding the fractional pointer value to an input of a lookup table. Ina further example, an output of the lookup table comprises N complexvalued numbers in a form s(0, . . . N−1); the method further comprising:performing a Fast Fourier Transform (FFT) on the selected whole samplesto create an FFT output, wherein the FFT has a length N and the FFToutput is a data array having values of a form d(0, . . . N−1). In astill further example, applying the complex time shift values comprisesmultiplying the complex valued numbers by respective output values ofthe data array in order to perform the sub-sample interpolation.

In one example, the fractional pointer value and the integer pointervalue are updated at each clock cycle of the coherent opticalcommunications receiver. In a further example, the method furthercomprises performing an inverse FFT on the interpolated signal samples.

In one example, the integer pointer value and the fractional pointervalue are derived from an output of a digital loop filter.

In one example, the integer pointer value defines a first sample of theselected whole samples.

In one example, the method is performed in a dynamic polarisationrecovery filter module. In a further example, the method furthercomprises modifying the first sample rate to a second sample rate,wherein modifying the first sample rate comprises: performing an FFT onthe selected whole samples; performing one or more dynamic filterfunctions; and performing an inverse FFT (IFFT), wherein the FFT and theIFFT have different lengths. In a still further example, the one or moredynamic filter functions includes multiplying, in a multiple in multipleout filter, the selected whole samples by respective frequency domaintap weight values.

In another form, an optical communications receiver is provided. Theoptical communications receiver comprises: an analogue-to-digitalconverter (ADC), the ADC being adapted to convert an analogue datastream derived from an optical data input to produce a digital inputsignal comprising a plurality of data samples at a first sample rate;and a digital signal processor adapted to: select whole samples from thedigital input signal in a time domain using an integer pointer value, toprovide selected whole samples; generate complex time shift values forthe selected whole samples using a fractional pointer value; and applythe complex time shift values in a frequency domain to the selectedwhole samples thereby causing a sub-sample interpolation.

In another form, one or more non-transitory computer readable storagemedia are provided. The one or more non-transitory computer readablestorage media are encoded with instructions that, when executed by aprocessor of a coherent optical communications receiver adapted toconvert an optical data input stream into a digital input signalcomprising a plurality of data samples at a first sample rate, cause theprocessor to: select whole samples from the digital input signal in atime domain using an integer pointer value, to provide selected wholesamples; generate complex time shift values for the selected wholesamples using a fractional pointer value; and apply the complex timeshift values in a frequency domain to the selected whole samples therebycausing a sub-sample interpolation.

What is claimed is:
 1. A method for digital resampling in a coherentoptical communications receiver, the coherent optical communicationsreceiver being adapted to convert an optical data input stream into adigital input signal comprising a plurality of data samples at a firstsample rate, the method comprising: selecting whole samples from thedigital input signal in a time domain based upon an integer pointervalue, to provide selected whole samples; generating complex time shiftvalues for the selected whole samples based upon a fractional pointervalue; and creating interpolated signal samples by applying the complextime shift values in a frequency domain to the selected whole samples toperform a sub-sample interpolation of the selected whole samples.
 2. Themethod of claim 1, where generating the complex time shift valuescomprises providing the fractional pointer value to an input of a lookuptable.
 3. The method of claim 2, wherein an output of the lookup tablecomprises N complex valued numbers in a form s(0, . . . N−1); the methodfurther comprising: performing a Fast Fourier Transform (FFT) on theselected whole samples to create an FFT output, wherein the FFT has alength N and the FFT output is a data array having values of a form d(0,. . . N−1).
 4. The method of claim 3, wherein applying the complex timeshift values comprises multiplying the complex valued numbers byrespective output values of the data array in order to perform thesub-sample interpolation.
 5. The method of claim 1, wherein thefractional pointer value and the integer pointer value are updated ateach clock cycle of the coherent optical communications receiver.
 6. Themethod of claim 5, wherein the method further comprises performing aninverse FFT on the interpolated signal samples.
 7. The method of claim1, wherein the integer pointer value and the fractional pointer valueare derived from an output of a digital loop filter.
 8. The method ofclaim 1, wherein the integer pointer value defines a first sample of theselected whole samples.
 9. The method of claim 1, wherein the method isperformed in a dynamic polarisation recovery filter module.
 10. Themethod of claim 9, further comprising modifying the first sample rate toa second sample rate, wherein modifying the first sample rate comprises:performing an FFT on the selected whole samples; performing one or moredynamic filter functions; and performing an inverse FFT (IFFT), whereinthe FFT and the IFFT have different lengths.
 11. The method of claim 10,wherein the one or more dynamic filter functions includes multiplying,in a multiple in multiple out filter, the selected whole samples byrespective frequency domain tap weight values.
 12. An opticalcommunications receiver, comprising: an analogue-to-digital converter(ADC), the ADC being adapted to convert an analogue data stream derivedfrom an optical data input to produce a digital input signal comprisinga plurality of data samples at a first sample rate; and a digital signalprocessor adapted to: select whole samples from the digital input signalin a time domain using an integer pointer value, to provide selectedwhole samples; generate complex time shift values for the selected wholesamples using a fractional pointer value; and apply the complex timeshift values in a frequency domain to the selected whole samples therebycausing a sub-sample interpolation.
 13. The optical communicationsreceiver of claim 12, wherein the digital signal processor comprises adynamic polarisation recovery filter module, and wherein the digitalsignal processor is arranged to generate the complex time shift valuesand apply the complex time shift values in the dynamic polarisationrecovery filter module.
 14. The optical communications receiver of claim12, wherein the digital signal processor is further adapted to: providethe fractional pointer value to an input of a lookup table.
 15. Theoptical communications receiver of claim 14, wherein an output of thelookup table comprises N complex valued numbers in a form s(0, . . .N−1), and wherein the digital signal processor is further adapted to:perform a Fast Fourier Transform (FFT) on the selected whole samples tocreate an FFT output, wherein the FFT has a length N and the FFT outputis a data array having values of a form d(0, . . . N−1).
 16. The opticalcommunications receiver of claim 15, wherein the digital signalprocessor is further adapted to: multiply the complex valued numbers byrespective output values of the data array, in order to perform thesub-sample interpolation.
 17. One or more non-transitory computerreadable storage media encoded with instructions that, when executed bya processor of a coherent optical communications receiver adapted toconvert an optical data input stream into a digital input signalcomprising a plurality of data samples at a first sample rate, cause theprocessor to: select whole samples from the digital input signal in atime domain using an integer pointer value, to provide selected wholesamples; generate complex time shift values for the selected wholesamples using a fractional pointer value; and apply the complex timeshift values in a frequency domain to the selected whole samples therebycausing a sub-sample interpolation.
 18. The one or more non-transitorycomputer readable storage media of claim 17, wherein the instructionsthat cause the processor to generate the complex time shift valuesinclude instructions that cause the processor to provide the fractionalpointer value to an input of a lookup table.
 19. The one or morenon-transitory computer readable storage media of claim 18, wherein anoutput of the lookup table comprises N complex valued numbers in a forms(0, . . . N−1), and wherein the instructions further cause theprocessor to: perform a Fast Fourier Transform (FFT) on the selectedwhole samples to create an FFT output, wherein the FFT has a length Nand the FFT output is a data array having values of a form d(0, . . .N−1).
 20. The one or more non-transitory computer readable storage mediaof claim 19, wherein the instructions that cause the processor to applythe complex time shift values include instructions that cause theprocessor to multiply the complex valued numbers by respective outputvalues of the data array, in order to perform the sub-sampleinterpolation.